MAX15022EVKIT+【PDF 17/28 页】KIT EVAL FOR MAX15022

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Dual, 4A/2A, 4MHz, Step-Down DC-DC

Regulator with Dual LDO Controllers

Below are equations that define the power modulator:

use aluminum electrolytic capacitors while for space-

sensitive applications, use low-ESR tantalum or multilay-

= = 4V/V

Gain MOD(DC) =

V AVIN

V RAMP

V AVIN

V AVIN

4

er ceramic chip (MLCC) capacitors at the output. The

higher switching frequencies of the MAX15022 allow the

use of MLCC as the primary filter capacitor(s).

2 π × L × C OUT × ? OUT

? R OUT + DCR ?

f LC =

1

? R + ESR ?

?

≈

1

2 π × L × C OUT

First, select the passive and active power components

that meet the application output ripple, component

size, and component cost requirements. Second,

choose the small-signal compensation components to

f ESR =

1

2 π × ESR × C OUT

achieve the desired closed-loop frequency response

and phase margin as outlined below.

Closed-Loop Response and Compensation

R OUT is the load resistance of the regulator, f LC is the

resonant break frequency of the filter, and f ESR is the

ESR zero of the output capacitor. See the Closed-Loop

Response and Compensation of Voltage-Mode

Regulators for more information on f LC and f ESR .

The switching frequency (f SW ) is programmable

between 500kHz and 4MHz. Typically, the crossover

frequency (f CO )—the frequency at which the system’s

closed-loop gain is equal to unity (crosses 0dB) —

should be set at or below one-tenth the switching fre-

quency (f SW /10) for stable closed-loop response.

The MAX15022 provides an internal voltage-mode error

amplifier with its inverting input and its output available to

the user for external frequency compensation. The flexi-

bility of external compensation for each controller offers

a wide selection of output filtering components, especial-

ly the output capacitor. For cost-sensitive applications,

of Voltage-Mode Regulators

The power modulator’s LC lowpass filter exhibits a vari-

ety of responses, dependent on the value of the L and

C and their parasitics. Higher resistive parasitics

reduce the Q of the circuit, reducing the peak gain and

phase of the system; however, efficiency is also

reduced under these circumstances.

One such response is shown in Figure 4a. In this exam-

ple, the ESR zero occurs relatively close to the filter’s

resonant break frequency, f LC . As a result, the power

modulator’s uncompensated crossover is approximate-

ly one third the desired crossover frequency, f CO . Note

also, the uncompensated rolloff through the 0dB plane

follows a single-pole, -20dB/decade slope and 90° of

phase lag. In this instance, the inherent phase margin

ensures a stable system; however, the gain-bandwidth

product is not optimized.

40

20

f LC

MAX15022 fig04a

90

45

80

60

< G EA

MAX15022 fig04b

180

135

0

|G MOD | ASYMPTOTE

|G MOD |

0

40

20

|G EA |

f LC

f CO

90

45

-20

< G MOD

f ESR

-45

0

0

-40

-60

-80

-90

-135

-180

-20

-40

-60

-80

< G MOD

f ESR

|G MOD |

-45

-90

-135

-180

10

100

1k

10k

100k

1M

10M

10

100

1k

10k

100k

1M

10M

FREQUENCY (Hz)

Figure 4a. Power Modulator Gain and Phase Response with

Lossy Bulk Output Capacitor(s) (Aluminum)

FREQUENCY (Hz)

Figure 4b. Power Modulator and Type II Compensator Gain and

Phase Response with Lossy Bulk Output Capacitor(s) (Aluminum)

______________________________________________________________________________________

17

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